The vacuum tube as amplifier

All tube types except diodes have the potential to work as amplifiers. A voltage change on the control grid will cause a corresponding change in the plate current, so basically, the tube is a voltage amplifier, but the input voltage signal is changed to a current signal. The trick when building an amplifier is then to utilize this current signal in a suitable way, either by using it directly or by changing it back to a voltage signal.

Early methods

The first tube amplifiers used transformer coupling. A transformer basically handles current, so by letting the plate current pass through a transformer, you utilize the current signal very well.

Another advantage of the transformer is that the winding ratio can be made to adapt the signal to the next stage; if the next is to be another amplifier tube, the transformer can be a step-up transformer, effectively giving up to 10 times extra gain (higher ratios tend to run into various practical problems). If the next “stage” is a loudspeaker, on the other hand, a step-down transformer can be used, matching the low impedance of the loudspeaker. Once radio technology came out of infancy, tubes became better and cheaper, so in AF amplifiers, the transformer coupling only lived on in the output stage, where low impedance loudspeakers would otherwise have been difficult to drive with the relatively high-impedanced vacuum tube. In RF amplifiers, transformers are simple and cheap and not only have they stayed the tube era out, but they live on in even the most modern solid state circuits. We will return RF amplifiers later.

Resistance coupling

Transformers for AF are bulky and expensive, so as the gain-factor of tubes became better, the resistor coupling became the method of choice for most AF amplifiers and even some RF amplifiers. A resistor is inserted in the plate circuit instead of the transformer. As the grid signal changes the plate current, obviously, the voltage drop over the plate resistor will change accordingly, giving raise to a voltage signal on the plate. This signal will be in opposite phase compared to the grid signal; when the grid signal voltage swings in the negative direction, the plate current will fall, thus the voltage drop over the plate resistor will be smaller, making the plate voltage rise, whereas a positive grid swing will increase the plate current and lower the plate voltage.

The plate signal will have a rather large DC component: It will swing around an average plate voltage, typically at around two thirds of the B+ voltage. To keep this voltage from entering the next stage, we use a coupling capacitor, or in older terminology, a blocking capacitor (because it blocks the DC, but lets the signal pass). Obviously, unpleasant side effects may occur if this capacitor fails and shorts out, especially if the next stage is a transformer coupled output stage: The output tube will get a positive bias making it draw an extreme amount of plate current, which may in turn overload and damage or destroy any or all of the following parts: Output tube, output transformer, rectifier tube(s), mains transformer. Some restorers of old radios for this reason prefer to replace all coupling capacitors as a preventive measure.

A look at the details

The values in the schematic are for a stage using Ѕ 12AT7 (ECC81). At 100V plate voltage, this tube will draw 3mA plate current with a grid bias of -1V, so to make the stage autobiasing, we need a cathode resistor R2 of 330 ohms. The decoupling capacitor C1 needs to have an impedance comparable to R2 at the low frequency limit, lets say at 20Hz, thats
1 / 2p *300 * 20 ~ 33MF (so this will be an electrolytic). The grid return resistor R1 is there because electrons leaving the cathode will occasionally hit the grid and if they are not led away from the grid, a negative charge will accumulate, eventually cutting off the plate current. This resistor just needs to be some large value, usually 1meg is used. The plate resistor R3: The setup with the bias used calls for the tube to have a plate voltage of 100V, and if we assume a B+ of 150V, this means that the voltage across R3 must be 50V, and at 3mA, that gives us 17k (standard value: 18k). A configuration like this, with about 1/3 of the B+ across the resistor will give a near optimum signal swing (when saturated, a vacuum tube has a considerable voltage drop over it, so a 100V swing is all we can hope for). Finally, the coupling capacitor C2 needs to have a low impedance compared to the input resistance of the next stage at the low frequency limit. Assuming 1meg and 20Hz, we get 0.01MF (easily within polyester range).

So, what gain can we expect from this stage? The conductance for the 12AT7 at this bias point is 5.5mA/V, so a 1V signal will ideally give a 5.5mA change in plate current. We might expect this to result in a 99V signal across the 18k, but life is not like that: A triode basically acts like a resistor with its value controlled by the grid. If this were an ideal resistor, we would expect it to be 100/3=33k, but the plate voltage has an influence on the resistance of the tube, so the dynamic impedance of the tube is only 16.5k (according to the data sheet). This impedance has to be calculated in parallel with the 18k plate resistor, giving 1/(1/18+1/16.5)=8.6k, so the actual voltage swing on the plate will be 8.6*3=26V, giving our stage a voltage gain of 26. Then there is the load of the next stage, but 1meg load will not make a world of difference.

Output stage

As I mentioned, one place the transformer coupling has stayed in use is in AF output stages. The signal impedance of even a powerful output tube is still in the kiloohms range, but loudspeakers typically have impedances of 4-8 ohms. The transformer is perfect for this, and at the same time provides for a safe isolation between the voltages inside the set and outside circuits like external speakers.

Above is a full diagram of a typical single-tube output stage cabable of delivering 5-6 watts of output power. Here, as mentioned, the transformer is virtually indispensable. A stage like this is biased for a quiescent current of half the full swing; this is called class A, a naming convention that has led some people to assume that such a stage is in some way better than a stage that runs in class B (discussed later in this article). We find another electrolytic here, again at the cathode resistor. If this capacitor shorts out, the stage will draw a lot of current, possibly with the effects described earlier, but the more common fault for an old electrolytic is that it dries out and looses most of its capacity. In this case, a negative feed-back occurs over the cathode resistor, which will reduce the amplification in the stage considerably, probably resulting in a low output volume.

A look at the details

Let’s start at the input: The grid return resistor R1 is empirically set to 560k (a big grid produces more return current), so the coupling capacitor C1 ends at 0.02MF. R2 is there to prevent high frequency oscillation (in spite being built for AF, this tube will work well into the shortwave band), the 1k value is again empirical. The cathode resistor R3 is 135ohms to give us a -6.5V bias at 48mA, this leaves C2 at 100MF. The output impedance of the stage with this bias is 5.2k. This value is equivalent to the load value we calculated for the triode stage (internal resistance in parallel with an imaginary plate resistor), and gives an output power of 5.7W, according to the data sheet. If the speaker impedance is 8ohms, the winding ratio of the transformer is the square root of the impedance ratio: 26.

With no signal, the full B+ (250V) will be at the plate since the DC resistance of the transformer is low (in the order of 200ohms). 5.7W in 5.2k is 172V RMS, 243V peak, so with a positive signal swing at the grid the plate voltage swings very low, but what with a negative grid signal? Well, the plate current will now start to fall from the 48mA, but the inductance of the transformer will make the plate voltage rise above B+ to nearly 500V, so the full voltage swing at the plate is a teoretical 500V PP! Whether the tube is actually able to swing down to a few volts is perhaps doubtfull, but th upward swing is a real thing, and now watch out: If the load is larger than 8ohms, as when the speaker is open, the voltage will simply rise till the power can be dissipated SOMEWHERE. I have seen the output transformer of a 2W amplifier arch to the iron core of the transformer. So contrary to what you might expect, underloading can be more damaging to this stage than overloading. An overload, like a 4ohms speaker, will just reduce the voltage swing at the plate, and we won’t get optimal power, but no damage will be done. If the amplifier is built with a high negative feed-back, the situation can be different.

More power

If we want more power, we could use a bigger tube, but the quiescent DC current in the class A stage poses some problems: To cope with the DC magnetization, the transformer has to have a relatively large core; in effect, we only use half the operating range of the transformer core, since it is only ever magnetized in one direction. Also, the power stage always draws power equivalent to full output, and this is really ineffective, since even if we run it at full volume, the average output power in music or speach will be considerably lower than the sound peaks. So a big amplifier will dissipate a lot of unneccessary heat. Instead, we can choose to run two tubes in a so-called push-pull stage, where each tube takes care of one half of the signal:

Right away, you might expect this stage to be able to handle twice the power than the one above, but because the two 6BQ5 tubes dont need to negotiate the high quiescent current, they are actually able to deliver up to 17W. Note the center tab on the output transformer: Not only is the quiescent current lower, but the bias current to the two tubes cancel each other out in the transformer. As a result, the transformer for our 17W amplifier need not be much larger physically than the one for the 5W class A stage. This class is called a class B stage, signifying that the quiescent current is at a level considerably lower than the mean signal bias. The tubes need two input signals, in opposite phase, so there has to be a driver stage that supplies this. There are several methods of doing this, they are discussed below.

Details

The obvious way to get a split-phase signal is of course to use a transformer:

This kind of construction can be seen in old sets, but again, transformers are relatively expensive, and also their frequency transfer charateristics are less than ideal, so there are good reasons to try and avoid them if possible. Interestingly, this configuration later had a revival: Transistor output stages used it extensively during the first 15 years or so of transistor radios. But for tube sets, other methods came in use. Some amplifiers use a simple phase splitter stage:

The cathode- and plate resistors are the same value, so equal signals, but in opposite phase, are found on cathode and plate. Note the voltage divider to the grid: Since the cathode resistor is far larger than in a normal self-biasing stage, the grid must be “lifted” to obtain a suitable bias. There is no overall gain in the stage, so a previous stage is usually needed to provide the neccessary gain. Also, as the output impedances of the two outputs are different, some imbalance can occur, especially at the extreme ends of the frequency range. This next driver construction, while slightly more complex, is better balanced:

The idea is simple; one stage is a normal amplifier, the next one is too, but a voltage divider at the input lowers the signal back to the level before the first stage, then it is amplified back up, and the phase is inverted. Note that the stages have no decoupling on the cathode resistors: This gives us precise control of the gain which will simply be equal to the ratio between the cathode resistor and the plate resistor.

Finally, the cheapie:

The idea is, of course, that if we feed signal to one output tube, there will be a nice inverted signal on its plate, which can then be fed to the other tube. This stage is inherently poorly balanced, because if tube no. 2 gets too much signal, the stage will oscillate, so to be on the safe side it has to be underbalanced. Obviously, the only advantage of this configuration is its simplicity, and it is probably quite rare, but does exist.

Bias considerations

As mentioned, the bias conditions for push-pull stages are normally called class B, but most tube amplifiers really run in what modern amplifier designers would classify as AB. The reason is that the bias is fairly high compared to class B solid state stages. This means that cross-over distortion is not too big a problem, but it is also a product of neccessity: Normally, the stage will be self-biasing, so to back the quiescent current way off, you would need a quite large cathode resistor, so the normal 15-30W radio or HI-FI output stage will run with a bias current of about 25% of max current. If bias is to be lowered to the 2-5% level of modern amplifiers, a negative supply is normally needed.

RF amplifiers

One type of RF amplifier found in nearly any receiver is the IF amplifier:

Obviously, this is really a transformer coupled stage. RF transformers are simpler and cheaper than their AF counterparts because they have fewer windings and the bulky iron core can be replaced by ferrite cores or nothing at all. Also, most RF amplifiers are tuned, so the transformer coil naturally becomes part of the tuned circuit. Note that a pentode is used: In a triode, the internal capacity between the grid and the plate acts as a parasitic feed-back path, and because of the phase transitions around the resonant frequency of the tuned circuits, the stage will have a strong tendency to oscillate. The screen-grid in the pentode disconnects this capacity, and makes the stage inherently stable.

At high frequencies, pentodes become a problem. Not only do they have a higher noise level than triodes, but the more complex tube system has to be physically larger, and this limits their performance at high frequencies. So here we are stuck with the triode, and the inherent instability if the triode stage must be overcome in another way. Take a look at this:

This configuration is called grounded grid, for obvious reasons. What controls the plate current is the cathode-grid voltage, so instead of holding the cathode at a fixed voltage (effectively grounding it) and feeding the signal to the grid, it is possible to ground the grid and feed the signal to the cathode. The grounded grid effectively screens the plate from the signal input, which is now the cathode, and thus the stability is vastly improved. The draw-back is that the stage has a low input impedance, but since we are using a transformer coupling this is not too big a problem. The impedance matching in the stage shown is achieved by feeding the stage from a tab on the input coil. The grounded grid configuration is widely used for RF stages at all frequencies above some 50mHz. For the real high frequencies, special triodes exist that can work effectively well into the gigahertz range.

The low input impedance makes the grounded grid configuration unfit for AF amplifiers, except for the special uses, where the low input impedance is desireable, like dynamic microphone inputs, etc. To get the best of both worlds, the cascode configuration can be used:

This set up, at the expence of an extra triode system, has the high input impedance of the grounded cathode stage, an excellent input-output separation, and a high gain, making it very well suited for tuned amplifier stages. It is rare in commercial receivers, not only because of its complexity, but also because it is difficult to AGC regulate.

Special amplifiers

We have looked now at two basic amplifier couplings, grounded cathode, and grounded grid. A third exists: Grounded plate:

As you can see, the plate is of course connected to B+, not to ground, but signal-wise, B+ and ground are the same (to signify this, one of the B+ filter or decoupling capacitors is shown), so what happens is that no signal occurs at the plate, instead the signal is taken from the cathode. The max voltage gain from this stage is 1, but it has a high current gain, or, in other words, it has a low output impedance. The output impedance is much lower than the cathode resistor, because a high negative feed-back exists within the stage. Any change in output voltage because of a load will change the cathode-grid voltage and be countered by a current change, so the output impedance equals the cathode resistor divided by the gain factor of the tube. The input impedance, on the other hand, is very high: The grid return resistor may be 1meg, but it is not referenced to ground, it is referenced to a point close to the cathode, where there is almost the same signal as on the input. Therefore the input impedance is equal to the grid resistor multiplied with the ratio between the two cathode resistors, if we assume it is in this case 10, it will give us an input impedance of 10megs. More sophisticated input back-off or bootstrap schemes can be employed taking the input impedance into hundreds of megs. The grounded plate, or cathode follower circuit is used whereever you need a high input impedance and/or a low output impedance.

We can couple a cathode to another and get this, the differential amplifier:

The circuit has here been drawn symmetrical, and what really happens is that the two tubes divide the current from the cathode resistor between them. At small signal levels, this current can be considered constant, which means that what really gets amplified is the difference between the two grids. Plate signals are in opposite phase. The signal between the two grids is called the differential mode signal, any signal that is the same on both grids is called common mode, and common mode signals are suppressed, how much depends on how well the stage is balanced and on how close the cathode current is to a constant current. The differential amplifier has high input impedances and can have medium to high gain. The circuit is mainly used in instrument circuits, but can also be seen as driver for push-pull output stages.

Finally, still mainly for instumentation purposes, DC coupling

These days, complementary solid state devices have made construction of DC coupled amplifiers a breeze, actually most solid state AF amplifiers are largely DC coupled, if only to save capacitors, but back in the tube era, getting that zero hertz lower frequency limit was a rather complex undertaking. Here is a simple two-stage DC amplifier:

There is really no getting around a B- supply here, and since the signal partly refers to B-, it has to be a good one, so often stabilizing tubes are used. I have assumed a -90V B- here, as it can easily be stabilized with a neon lamp. The trick about it all is the voltage divider, which gets us back to an appropriate level for the second stage. This of course costs us some gain, as the signal is also divided. The capacitors around the voltage dividers are there to compensate for various stray capacities, most prominently the cathode-grid capacity of the second tube, which would otherwise form a low-pass filter together with the resistirs, and lower the upper frequency limit considerably. The capacitors are quite small, in the MMF range and basically inversely proportional to the resistors, but the exact values will have to be determined empirically, because of the influence of stray capacities. Properly built, this amplifier will be linear from DC to over a megahertz. The method of transferring a DC signal by means of a voltage divider to a negative supply can be applied to most amplifier configurations.

There are quite a few interesting details: We are using pentodes, because they have better gain potential than triodes. The first tube is auto-biasing, this allows us to have the grid at zero potential. The cathode resistor is of course not decoupled, since we cannot decouple for DC, anyway we want control over the gain and we want the linearity from the negative feed-back it offers. The next tube also has a cathode resistor although we could easily bias that via the voltage divider that transfers the signal, but we also want the negative feed-back here. The screen grids need 120V for the bias point we are using; to get that from 250V, we would normally just fit a 56k resistor, which would get us close enough with the 2.4mA drawn by the grid, but as the screen grid cannot be decoupled, our gain would be reduced, so we need to add some stabilizing. A really expensive amplifier would have a separate B+ for this, but here we just use a voltage divider with some extra current running through it to get a more stable screen grid voltage. In a real application, some bias adjustment would be needed to trim away the DC offset that will invariably occur. This amplifier has a total gain in excess of 250, and some negative feed-back might be employed to improve linearity and frequency range.

Guitar amplifiers should always be tested before purchase (most music stores will let you play around a little) to make sure they are compatible with your guitar and produce a good clear sound. If there are any doubts in the music store, don’t’ buy it! Some amps will sound better with certain brands of guitar; bring yours in with you to be extra sure of your purchase.

Finally, guitar amps are all about volume. What’s the point of playing if no one can hear you? As with sound quality, try out a high volume on an amplifier before you commit to buying it. If the sound goes crackly, it’s not right for you. If you were to go on stage at any point, you’d need to push the volume right up to be heard over the rest of the band, and crackling speakers are out. As a simple rule, test out an amplifier before you buy it. If it gives you special effects, controlled sound and good volume, you’ve got yourself a great amp.

Six JAN (joined army navy) tubes, after some 40 years of rest, finally make music…

I started this project with the idea to build a simple single-ended tube amplifier. I didn’t have any experience with the construction of tube amps so I was looking for a beginner’s project. Such a project should be simple (for easy debugging) and it should also be cheap. Nobody likes to blow up a component of 300 dollars by a small lapse of attention…

As far as simplicity is concerned, nothing beats the “Darling” series of amplifiers which are described on the website of New Jersey resident Bob Danielak 1. They employ power triodes type 1626 which can be obtained cheaply from various sources (I bought several for 3 US dollars apiece). Old readers of this page may remember that the 1626 was employed in various transmitters for the American airforce, e.g. the AN/ARC-5 series (BC-457, 458, 459 and 696). Since they were used in military airplanes, 1626s are rugged valves which can stand shocks, vibration and electrical insults. Once I had excessive power dissipation in two 1626s because of a wiring error. The plates became fiery red and blazingly hot and remained in this condition for two minutes before I discovered my mistake. Yet these tubes still sound and measure OK. Apparently, Darling amplifiers make excellent projects for dummies like me!

I decided to make Jeremy Epstein’s DC-coupled variant of the Darling amp with two parallelled 1626s per channel 2. The schematic is shown below:

For my first try, I followed Jeremy’s schematic to the letter, although I used a GZ34 rectifier in stead of a 5U4G. The Hammond 125E output transformers, 1.5 H choke plus the 8532 and 1626 tubes were from Antique Electronic Supply in Tempe, AZ. A NOS power transformer (Prova) supplied 2 x 345, 5 and 6.3 Volts. I acquired this tranny and the GZ34 rectifier from Frits Meuris Electronics (Sittard, Holland). The 12.6 Volts for the 1626 heaters came from an obscure transformer from my junkbox which I bought for one Dutch guilder from Conrad Electronic.

I built the amp in the old style, hardwired on an aluminum chassis in a wooden frame with all tubes, chokes and trannies on top and all electrical wires below. Special parts were not used. I took what was in my junkbox (carbon resistors and styroflex capacitors from the former DDR, electrolytics from Czechia, wirewound resistors from Vitrohm in Germany). After I had corrected my wiring error, all voltages were within a few percent of Jeremy’s values and 1 kHz square waves looked OK. No trace of microphonics. So on for the listening tests!

After a burn-in period of a few hours, I listened to live recordings from twentieth-century classical music (Schцnberg Ensemble), using the multimedia speakers which are described elsewhere on this website. The amp sounded wonderful, as if you were in a very good seat in the concert hall. I was really excited about the “SE magic”. But after a while I became disappointed. The amp sounded excellent with some programme material (chamber music, small combos), but it didn’t sound good at all with large orchestra, grand piano, church organ, or big drums. The presentation seemed then “anaemic”. Especially piano recordings sounded terrible. Measurements indicated what was wrong:

The frequency response of this version of the amplifier was 75Hz…14 kHz (-3 dB). With orchestral music and grand piano, I was missing the lower two octaves. For Jeremy this didn’t cause problems, for he employs his own Darling as the treble amp in a bi-amped system. But as a stand-alone amp, the results were not satisfactory.

Jeremy thought the cheap Hammond transformers were limiting the bandwidth. So I decided to try more expensive output trannies. I replaced the Hammonds with Lundahl LL1664s from Sweden which I purchased from Aqua Blue in Belgium. The (almost fivefold) difference in price certainly resulted in improved specs (see below).

With the Lundahls, the frequency response was flat from 8 Hz…55 kHz (-3 dB). This was a major improvement! Grand piano now had weight and sounded like the real thing. Sibilants were somewhat emphasized in the Hammond 125E version, but sounded natural with the Lundahl LL1664.

So I was very happy for a while. But the perfectionist bug hit me once again. IMHO, the amp still had a major flaw. There was audible hum in the output. With the multimedia speakers, the hum was not objectionable during the reproduction of music. But in silent passages, it became annoying. Additional measurements indicated that the signal-to-hum ratio was only -55 dB with shorted inputs!

First I thought that I had made a grounding error, but this proved not to be the case. Then I thought: maybe I should use DC heating. However, there was no improvement when I hooked up the 8532 heaters to a DC power supply. I noticed that the hum had a frequency of 100 Hz rather than 50 Hz. I concluded that it originated from the B+ supply.

After running some simulations on Ben Duncan’s PSU designer program, I decided to modify the B+ circuit. After the GZ34, I maintained the 47 uF electrolytic capacitor. The stereo channels are connected to this cap via individual LC filters (10 H chokes and 200 µF/500V electrolytics). This modification resulted in a 29 dB improvement of the signal-to-noise ratio. The capacitive load to the GZ34 remains within safe limits. And hum and noise are now completely inaudible.

Specifications of our modified Darling:

Output power: 2 x 1.5 Watts in 8 Ohms

Voltage gain: 8.6 x

Input sensitivity: 400 mV

Frequency response: 8 Hz…55 kHz (-3 dB)

Signal-to-noise ratio (guesstimated): -84 dB (related to 1.5 W output)

This is an excellent amp. It sounds lively, agile, and natural. Makes you tap your feet and sing along with the music.

Verdict after two years of use: We listen daily to this amp on the large Jericho horns, so you can guess that we are pleased with its sound. However, the NOS power transformer (Prova) suddenly broke down (catastrophic damage to the insulation of the B+ windings) and we had to resurrect our killed Darling, using a custom-made transformer (purchased from Kent Electronics). We now used a metal chassis (Hammond) and we replaced the 1000µF 250V electrolytics with 47µF 400V polypropylene capacitors (Audyn-Cap, Intertechnik). This slightly improved the sonics. With the electrolytics, the sound could occasionally be somewhat “hard”, with the foil caps this never happens, not even in difficult passages. This tube amp sounds sweeter than any of my solid-state products. Finally, we added volume controls (100k log). Now, the amp can be driven directly by line-level sources (CD players, FM tuners). A preamp is no longer necessary.

Finally, a picture:

Darling (front view). A protective cover was mounted over the transformer after the testing phase.

References

¹ Bob Danielak, “Darling” and “DC Darling” SE 1626 amps”

² Jeremy Epstein, “My experiments with Bob Danielak’s ‘darling’ 1626 amplifiers”

If you need a demonstration of retuning’s musical impact read this paragraph, stop, and do the following. Pick about ten bars of a familiar record and play it a few times. (use a record you don’t like if you’re concerned that quick successive replays will hurt.) Become familiar with the sound (female voice is best). Now change the tracking force. No, don’t get out the gauges–just add or delete what might be a tenth or two of a gram. Hear the difference?–whether for better or worse. That’s one small change in a series of small changes is available.

Being persnickety helps you get the most from your LPs because you’re operating on such a minute scale. The grooves of a record are a few thousandths of an inch wide. Depending on the loudness at which the system is being played, you can usually hear down about 60+ dB, which means you’re hearing groove displacements of the order of a few millionths. (That’s like splitting a hair into one thousand pieces.) Every bit of motion or vibration allowed at this level can be heard through your speakers–greatly amplified.

What follows is a basic primer for table setup. To be more comprehensive here is impractical, if not impossible–spelling out how to optimize one product alone would take up pages. Instead, this gives the basic rationales for each procedure, along with some guidance as to what to do in each case. It offers a starting point for your own explorations or at least introduces you to the essentials of setup and fine-tuning, which may then encourage you to seek out someone familiar with the particularities of your own table. If you feel you’re a fumblefingers, don’t proceed. (You could cause some expensive damage.) Find instead a local expert to perform the magic. (Just be sure this person is an expert, is familiar with your particular table, and has set them up before.) This primer does not supersede the owner’s manual, which should be your primary guide.

Another factor to consider: If your cartridge is getting on in life,much of the following may not have the sonic impact it should. There is even a small chance that a worn stylus is damaging your records. Cartridges are one of the most difficult (and most expensive) purchasing decisions in hifi because it is impossible to get them on loan. As an interim measure (before chancing big money on a major “name” cartridge), you might investigate one of the highly-rated inexpensive units. On the other hand, don’t get hooked into the cartridge-of-the-month syndrome. Older, toprated cartridges with thousands of hours use can sound nearly as good as the best of today.

At various steps along the way in this retuning, your system may not sound as sweetly musical as at other times. Beware of thinking you have made the wrong adjustment. Many times, you will make a technical improvement which will reveal a previously underlying nasty sound. Try and fix the nasty sound, don’t just go back to the previous setup. If it sounds cleaner in the very bottom, and less “wooly,” you have probably improved things. On the other hand, if nothing has changed except that it now sounds “nasty,” then you probably erred in the adjustment.

Turntable Adjustments and Maintenance Support and Vibration

The first area to examine is the foundation of the entire turntable system, whether shelf or stand. No matter how good the table’s suspension, some vibration will get through and muddy the sound from the bottom end to the midrange. Setting up the foundation to convey as little vibration as possible will help minimize the muddying. This is even more important for a turntable with no suspension.

If you can feel any motion of the foundation by lightly touching it with your finger tips while playing music, this is degrading your sound dramatically. To get a hint of just how great the effect is, listen to it through a stethoscope placed on the table or on its support. Or place a glass of water on the support and watch the water’s surface while playing music or walking around–this is a simple and graphic way to see how much acoustic and mechanical vibration is reaching your system. Remember that your hi-fi is trying to reproduce groove modulations as small as a few millionths of an inch–about 1/1000th the thickness of the hair on your head. Not an easy task within this vibrating environment.

There are several steps you can take to minimize motion induced by the playing of the system as well as motion present in the environment. The record player stand must be on a stable surface–flexing floor boards do not make a secure base. If you have the option, mount your table support on a masonry wall or floor–remember the table can be either inside or outside your listening room. If your floor is wood, perhaps you can stiffen it from beneath, for example by bracing a strut between basement floor and turntable stand. If you cannot cure floor-flex, mount your table on a rigid wall.

Be aware that moving your table to a more stable location may result in an apparent decrease in bass. Since the more stable location has less vibration, the support vibrates less and therefore feeds less back into the system. This is not a mistake. You have indeed improved matters; you’ve just altered the apparent subjective frequency response. Don’t reverse the move; correct the balance. To rebalance the system,you can try moving the speakers, or improve cartridge alignment, or play with room changes or even component changes.

Next, turn your attention to the stand or mount itself. All universal stands have some flat plate or bars which form the top and on which the turntable rests–this itself will vibrate harmfully (the weak point of universal record player stands). The thicker (read: stiffer) this is, and the more inert, the better the sound–and standard units are none too stiff. Don’t wimp-out on the replacement. Get something very heavy (at least 25 pounds, preferably much more) and thick (over three inches). The stand should be spiked to the floor–nearly all come this way.

(Tip: Experiment with the sonic differences of placing Sorbothane vs. spikes between table and stand. The Sorbothane partly isolates, while the spikes tighten the connection.)

More About Vibration

Turntable screws may loosen over time, allowing more parasitic resonances to occur. Be aware that overtightening can warp the mating surfaces and make matters worse. Then use your noodle, look at the size of the screw, and snug it up. This goes for all screws used to hold anything together, be it cartridge-to-arm, or wire-to-box. A few tables are designed to need tuning of some elements by fastener tightness; in these cases, follow the manufacturer’s recommendations.

(Tip: Consider adding damping material between two contacting pieces to dampen vibration, especially over big flat areas. The idea is not to have a squishy interface but to fill in the very small gaps left through manufacturing tolerances. Take apart the pieces, add a very very small amount of Blu-Tac [now available here] or any other non-hardening putty, then reassemble and tighten down until the parts are solidly back in contact. Where there are accurately machined, ground, or lapped surfaces in contact, use some sort of inert grease such as an industrial vacuum grease.)

Levelness

When a turntable goes out of level, generally the platter bearing’s performance and the arm’s dynamics, specifically anti-skate, are negatively affected. Because the platter bearing is round in a round sleeve, unlevelness alters how the bearing floats the bushing (except cases like the Well Tempered and the Versa Dynamics); the better the bearing, the less the effect. Sonic problems due to being out of level are greatest with a pivoting arm; least with a linear tracking arm under motor control.

So be sure your table’s platter and tonearm mounting board are on the level. Don’t just eyeball it–use an accurate level. If the platter is out of level,adjust the suspension (in the case of a suspended subchassis design). If the arm board is not level (which means the arm pivot is not vertical), either return it to your dealer for repair or re-level it yourself by shimming between the mounting board and its support.

Platter Bearing

About the only thing you can do here is to replace (or top up) the bearing oil. Follow the manufacturer’s recommendation as to how often and with what. Lift out the platter, sop up the old oil with a lint-free cloth (or suck it out with a clean eyedropper or syringe), then pour in the new, being careful not to make a mess by overfilling the well. (The shaft of the bearing takes up most of the room in the bearing well.)

(Tip: Most oil bearings will be improved sonically by a stiffer [higher viscosity] oil. However, if the motor drive system is not very robust, this stiffer oil could slow the system down. Most manufacturers sell their own high viscosity oil; on the other hand, experimentation can be fun.)

Drive Belt

Some belts are meant to be talcum-powdered, some to be slick; some are meant to be soft-faced (matte rather than shiny), some to be clean. Check with the manufacturer about the need and method for cleaning to maintain proper traction. Some tables, because of their motors, require slippage to start up and slow down smoothly so belts on these most likely are talced. Years of slippage will wear the talc off and then start to buff the belt shiny. In a case like that, replace the belt with a manufacturer’s original.

Platter speed is sometimes controlled by what part of the pulley the belt rides on, so be sure to get this right. Belts can be finicky about just where they ride on platter and pulley–be patient. Everything that is on the table when playing a record–platter, mat, record, clamp–must also be on the table when you install or adjust the belt on a suspended subchassis table. On a two-part platter, place the outer ring upside down on the inner and lay everything else on top. This will accurately weight the suspension while allowing you to view the belt on the pulleys.

Suspension

There’s not much you can do in the way of adjusting a non-suspension table, except to regard its entire support system as being a part of the table’s suspension. Refer back to that section and consider even more strongly how to improve the foundation’s vibration protection.

Suspension designs are all a little different so to adjust your suspended table, follow the manufacturer’s instructions. As suggested earlier, if you aren’t familiar with working on your table, find someone who is an expert at it. Tweaks peculiar to each record player which can significantly benefit the sound are discovered by users and fine-tuners over time.

If, you adjust the springs, you need to gain access to the underside of the table, raise it up on four soda cans. Everything that is on the table when you play a record–platter, platter mat, record clamp, and record (use one you don’t care about)–must also be on it when you tune the springs so the weight (and therefore position) is accurate.

Generally, you rotate the entire spring to adjust the suspension’s up and down motion, or rotate the nut at one end of the spring to adjust height and levelness.
Make small incremental alterations and check the results each time. The platter should float exactly the same distance about the plinth all around and the tonearm board must remain horizontal with the plinth. Pushing at the center of gravity of the suspended part of the table should, with most designs, cause the suspended part to move straight up and down very freely and not transition to sideways or rotational motion before the motion subsides. Keep adjusting until you can achieve this condition.

Arm Adjustments

The arm is pretty much maintenance- and adjustment-free. Snug up the arm mounting screws.Check, on a typical pivoting arm, that the bearings are sound: grasp the headshell and very, very gently attempt to move the arm back and forth along the length of the tube and rotationally. If you can feel any free play at the headshell,you’ve got a serious problem–get it fixed or replaced. Exceptions are the Well-Tempered or unipivot arms where by doing this you are causing it to ride up off the pivot.

If you have a viscous damping trough, be sure it contains the correct amount of damping fluid; it doesn’t evaporate but it does migrate. If there is dust and lint in there, clean it out and refill with the manufacturer’s damping material. Also, in the case of a variable paddle system like the SMEs, reassess whether you are using the correct paddle. Too much damping will make the sound tight, but will lose lots of fine detail; too little and the sound will be open and relaxed but also more hazy and smeary.

(Tip: To minimize arm tube resonances [which can add much high frequency hardness to the sound], damp the arm tube with a brushed-on coating of liquid latex [thin cosmetic grade for theatrical use is good], or heatshrink tubing, or a non-hardening putty like Blu-Tac.)

You’re trying to align the cartridge stylus with the record groove in as close a replication as possible to how the cutting stylus originally cut the record groove. You’re trying to untrace with your playback stylus what was traced with the cutting stylus–the closer the alignment of the one mirrors the alignment of the original,the more accurately it can read the grooves. Alignment needs to be optimized in three different planes. However, it cannot be equally perfect in each of the three, so it must be optimized for an overall best balance or compromise. Final adjustment must always be done by ear and over an extended period of listening time. Just to add to the complexity,each record is cut a little differently. Here again, optimize for an overall balance of good sound over a wide range of records (or adjust VTA for each record, which some people do if they have an easy VTA adjustment on their arm).

The three alignment planes are as follows. (Please note that it is the stylus, not the cartridge, that is being aligned.) First, viewed from above, the cartridge’s arcing movement across the record must maintain the stylus in the same relation to the groove as that of the cutting stylus’s straight-line tracking; this is Lateral Tracking Angle, or Tangency. Viewed from head on, the stylus must be perpendicular in the groove so as not to favor one groove wall, and therefore one channel, over the other wall/channel; this is Azimuth. Viewed from the side, the stylus must sit correctly in the groove, at the same angle as the original cutter; this is Vertical Tracking/Stylus Rake Angle. (VTA, however, varies from record to record. Therefore, this alignment must be set by ear, even more than is the case with the other adjustments.)

Also confirm that the distance from the center of the arm pillar (the upright post) to the spindle (usually fixed by the arm mounting board) is correct as this will affect the ability to achieve the tangency adjustments. This “L dimension varies with every pivoted arm–check your manual or with the manufacturer.
Essential tools are an alignment gauge, a tracking force guage, a record you don’t care about as accidents can happen, a strong light you can focus where needed, and screwdriver. Small needle-nose pliers and a magnifying glass or plastic magnifying card can be handy. It’s very difficult to make an accurate alignment gauge (do not relay on the accuracy of the gauge that comes with every arm), so get a good one. If it doesn’t snugly fit over the spindle, throw it out and get another.

Make sure that the arm’s wires, wire clips, and solder joints are in very good condition. At minimum, clean the contact between cartridge pins and wire clips by removing and replacing each clip. Holding the clips with needle-nose pliers can make this easier, but be careful that you don’t strain the wires where they join the clip. Check your cartridge mounting screws. Because these must be snugged tight, plastic screws are no good. Aluminum, brass, or stainless steel crews, provided they are new and the threads aren’t distorted, are fine. Allen head screws are great because the Allen wrenches used on them provide excellent leverage. To exert sufficient tightening force on a slotted head screw, you need a screwdriver with at least a 3/4″ diameter handle–jeweler’s screwdrivers just don’t do it.

To Get Started

Tape the platter securely to the plinth. If it can rotate during setup, your alignment measurements won’t be accurate. Just be sure taping does not alter its height or levelness. If this is not already done, mount the cartridge in the headshell and the headshell on the tonearm. The headshell screws should be finger-tightened just enough that the cartridge cannot fall off but is still loose enough that the cartridge is easily moved around. Work whenever possible with the stylus’s safety cap in place.

Set tracking force at nominal, then do the tangency alignment procedures, then the azimuth. Do not deviate from this sequence as each step affects the subsequent one–change the order and the setup will be wrong.

Tracking Force

This adjustment on the tonearm counterbalances the weight of arm and cartridge. At this point, use your tracking force gauge and setting tracking force according to your cartridge instructions–final adjustment will be done later by ear. If you do not have a tracking force gauge, but the arm does have a calibrated counterweight,defeat the arm’s anti-skate mechanism or set it to zero. Set the counterweight so the arm is level and balanced. Be very careful of the unprotected stylus–you cannot do this with its safety cap in place. Once the arm is balanced, lock it in its cradle and, using the calibrated counterweight, set the tracking force according to your cartridge’s recommended weight.

Tangency Alignment

Follow the instructions in your owner’s manual and those provided with your alignment gauge–different gauges use slightly different methods. As you square up the cartridge body with the gauge’s markings, be sure that the cartridge sides are square or your alignment will be wrong. When all adjustments are correct, carefully snug down the cartridge mounting screws. Keeping a firm grip on cartridge and headshell together so nothing shifts, delicately tighten each screw down a turn or so, then repeat until tight. Snugging down one screw all the way before tightening the others is almost certain to twist the cartridge out of alignment. However careful you’ve been, always check the alignment again after tightening.

Azimuth

The old mirror alignment technique for azimuth may work fine for some cartridges, but a hand-made moving coil cartridge cannot control this alignment well enough. The stylus may be several degrees away from perpendicular to the top of the cartridge.
There are two accurate ways to adjust azimuth. One is using your ears for the best sound. Rotate the cartridge in tiny, tiny increments, in different directions, getting a feel for the area where you get greatest stage width, depth, and so forth. The drawback to this approach is that, until you develop a good deal of experience with it, you can be confused by the changes in sound, so be patient and work carefully–it will give you the best results. The only remaining foolproof method requires using a voltmeter and a test record. Set the azimuth so that crosstalk at 1,000 Hz is the same for both channels.

Vertical Tracking Angle

Unless your tonearm has a special VTA adjuster, adjusting arm height can be a major nuisance, and particularly so if the arm pillar is held at a selected height only by a set screw. In these designs, altering height means releasing the setscrew, which usually results in the arm pillar dropping precipitously, leaving you in the dark about the original point from which you are now trying to add or decrease height. (I speak from bitter experience.) Jam the gap between pillar neck and collar with business cards so the pillar cannot fall when released or find/make a block that fits between the arm mount and the underside of the arm structure. See your tonearm manual for its recommendations on adjusting arm pillar height.

The best approach is to tune-in VTA gradually by listening to music. You know the arm needs to be lowered at the arm pillar when the overall sound is hard and bright, with thin bass or no deep bass, edgy highs, and harsh midrange (of course, this could also be tracking force which is too light). Distortion obscures low level details between the musical; notes so dynamic range is reduced. Transient attacks may be too sharp. Raise the arm when the sound is dull and damped, the highs rolled off, the lows muddy and lacking definition, and transient attacks are dull. Mind you, this sounds an awful lot like the effects of changes in tracking force (too light is edgy, too heavy is heavy and dull). They are different sounding but hard to explain.

Start with the arm a little low and very gradually raise it, first to where it is parallel to the record, and then so the back of the cartridge is tilting up. Keep track of your settings so you can return to the one you like best where everything snaps into focus. The range of adjustments can be quite broad, as much as 3/4″ or even more (at the arm pivot). Play with the full range so you know what it sounds like and don’t be diffident.

Antiskate Force (pivoting arms only)

This applies an opposing, balancing force to the natural inward drag of a pivoting arm while playing. Left uncontrolled, the stylus would push up against the inner groove wall, causing distortion both from mistracking and a cantilever skewed in relation to the cartridge generator. To set, lower the stylus down near the label of a record with a wide run-out to it. Increase antiskate until the arm starts to slowly drift outward, away from the label. Again, this should be finalized by ear as you listen to music. If image placement is a little off-center, or if things don’t seem to be locked in solidly, experiment with antiskate. Also, watch the stylus when you set it into a groove. Does it move to the right or left relative to the cartridge body? This indicates too much or too little antiskating.

Fine Tuning

You’ve got three adjustments roughed in at this point: tracking force, VTA, and azimuth. It’s a matter of reiteration to optimize the sound. The change in sound with each of these individual adjustments can be similar. It’s therefore necessary,in optimizing all three, to experimentally move from one type of adjustments to the next,then to the next, in order to balance the optimization for all three. Listen to female voice as you work; got for the maximum vocal character and a tactile sense of a person.
You want to start to deviate from the cartridge’s recommended tracking force by small increments. You are trying to put the electromagnetic system in its most linear position. Too much tracking force and you’re moving the coils (or moving magnet) out of the center position of their range. A tiny increment may be 100ths of a gram or less; but try as much as 0.2 of a gram deviation above and below the manufacturer’s basic recommendation in your experiments. Don’t worry about record damage from heavy tracking; most record damage is actually caused by mistracking in the middle-to-high frequencies with too little tracking force rather than with too heavy tracking. (Besides, 0.2 gram over is not heavy tracking at all.) That’s providing that the stylus is reasonably clean and in good condition. If you’re getting mistracking at the low (lightest) end of the range and yet the low range is generally sounding the best (and on moderate signals, not The 1812 Overture), then chances are you have either a dirty stylus, a bad record, an accumulation of crud in your cartridge, or a cartridge that’s getting old.
Changes in tracking force can change how you want VTA and azimuth adjusted. If azimuth was initially adjusted by ear, experiment with it. However, if it was set with instrumentation, leave it be and instead play around with VTA and tracking force. I sometimes think of this process as being a little like tightening down a series of screws–you do each a turn or two at a time and keep going round and round until you’ve got them all evenly snugged down and the surfaces mated without warping. Keep on patiently adjusting until you recognize that the sound is right and just locks into place.

(Tip: Some people find that degaussing [Fluxbuster] of a moving coil cartridge is recommended as often as every day, even if the cartridge hasn’t been used.)

OK, you’re now basically done. Final-most tuning will take days or weeks and is a matter of listening to the system in a relaxed way. Eventually little aspects of sound from one record to another will begin to annoy out of the overall good sound.This may range from too light tracking force to VTA. (Most good cartridges are temperature sensitive. When too warm, they get muddy, when too cold, they can get strident. Keep up with this as the seasons change.) Excluding people who adjust VTA with every record, most people will be very happy with a VTA position which is a good overall compromise for the records that are their favorites. So turn on the system, let it warm up, sit back and relax, and enjoy listening to the music even as you keep one ear peeled for further refinements.

One last, and important, word on stylus cleaning. There are multiple recommended stylus cleaning procedures, ranging from ultrasonics, manually brushing, even using sandpaper, and with various solutions-anything from the proprietary Freon-based solutions to just alcohol or alcohol and water, as in record cleaning solutions. These can have an effect on the shape and condition of contaminants left on the stylus. With some modern cartridges with very fine-line styli, it might be necessary to clean the stylus once per LP side. Different methods of cleaning may result in different sound a more or less frequent need for cleaning. Experiment with different methods–some sort of cleaning is essential.

(c) absolute sound

So I sat down with a calculator and the tube datasheets, and came up with some circuit values. But I forgot about the loading of the output stage, so the driver stage didn’t work correctly the first time I tested it. The driver stage clipped before the output stage… But I recalculated the driver stage, and the circuit then worked. I built my “prototype” on a piece of wood. The lethal result can be seen below.

The Hammond transformer didn’t have a heater winding for the 2A3, but it did have a center-tapped 5V rectifier winding, so I used half of that for the 2A3 heater. But then I didn’t have a heater winding for the rectifier… So I found a nice “junk” tube solution - the 6BY5GA! A dual rectifier or damper tube with a 6.3V heater. The 6BY5GA has a very low voltage drop, but it is not meant for use with large input capacitors. So I used a 5H choke for a choke input filter. The resulting B+ voltage is about 285-290V from a 600Vct winding!

The amp sounds great, even with the small output transformer. The dual 6SN7 input/driver stage makes the amp very sensitive - a bit too sensitive probably… I have replaced the Chinese 2A3 with a Sovtek. Both tubes sound good, but the Sovtek is a bit more HiFi’ish. I have bought some better output transformers, but I haven’t installed them yet. When something sounds good I don’t mess too much with it. I also have a Hammond chassis waiting to make the amp a bit more cat safe…

The schematic can be seen here:

I seldom design SE amp more than 2 stages, but my client request for higher gain. 6SL7 can give enough gain with only 2 stages but less current, I prefer higher current, and the sound quality of 6SN7 but less gain, that’s why I’m using 3 stages in this design. The 2A3 working at plate current of 60mA with cathode-plate voltage of 300V. The EH’s 2A3 can handle this 18W of dissipation. I had tried to change it to chinese 2A3 with better sound quality but it can’t take that high dissipation, need to reduce to 14W. Since my client love the EH’s sound, I leave the EH tubes on. I use DC filament supply from 5V ac, using a bridge rectifier and a 0.25 ohm resistor to get the 2.5V DC. If I build 2A3 for myself, I will use AC as filament supply, and use tube rectifier and choke to supply B+, that will sound even better, I think.

Nothing much to talk about the sound quality. As expected, 2A3 always put a grin on my face every time I I had build one and listen to it for the first time. So far I haven’t hear anyone says that he or she don’t like the sound of 2A3 SE amp. Tell me which 2A3 SE amp sound bad if you could find one……….

I’m designing another 2A3 SE amplifier (for my own use) that using 717A door knob tube, and will put it on here when finished. Now I’m still trying to find out the characteristic of 717A tube, and optimized it to drive 2A3. I can’t get complete data sheet for 717A, if anyone of you found the data sheet, I would like to have a copy!

The schematic of this 2A3 will be on soon, I’m quite lazy to draw it using any design software, I will draw it manually and scan as jpg file. Stay tuned for the schematics and the new 717A driven version.

The simple but messy layout, I build it without a schematic in hand, The 2 big rectifier is DC for filaments. I don’t like using DC for 2A3’s filaments. Again, customer’s request that I must follow. I really NOT enjoy building this 2A3! can’t follow what I like ,such as rectifier tubes, AC 2.5V filaments, oil caps etc .

Single-Ended Triode (a.k.a SET) amplifier producing 1…1.5W into 8 ohm load. Uses two Russian deflector combo 6f5p tubes wired as triode.

Introduction

6F5P is Russian TV deflection triode/pentode combo tube, somewhat like ECL85/6GV8. If the pentode section is connected in triode mode, the tube provides a great playground for experiments with 2-stage SET designs on a “dime”. It is no joke, however. The plate currves are very linear, reminding 45 or 2A3, input capacitance is low and the plate resistance is < 800 Ohm.

The schematic is as studied with SPICE in [4], with the following changes:

(1) output bias resistors were adjusted for specific tubes, for best results, assuming the bias voltage ~30 V and current 35-40mA; (2) bypass caps are 47uF; (3) instead of 2k driver stage autobias resistors were used series connected 1.6k constant resistor and 2.2k pot in rheostat wiring - one leg to the 1.6 resistor, two other legs - shorted and connected to to the ground (4) driver stage load is 2 resistors in series, 47k and 30k, with a switch shorting the 30k resistor. (5) power stage bypass caps can be disconnected with a switch.

The power supply uses ultra fast diodes. RC filters a separate for each channel.

For those preferring tube rectifiers - it makes sense to organize the supply around the theme of paper and oil capacitors and filter chokes:

This supply needs a ~600V CT transformer with a winding for the heater of EZ80. EZ81 (6CA4) can be used too; it is more robust but needs more heater power. A good choise is 6X5. 5AR4/GZ34 can be used, too (different pinout filament supply: 5V, 3A), but it is an overshoot. In case of EZ80 and 81 heater voltage is 6.3V and may be taken from the same coil that feeds 6F5Ps, although it is preferable to use a separate coil. If the same coil is used, pins 3 and 4 must be disconnected. One 6X4 per channel can be used as well.
The chokes are 5…9H. An excellent choice is to use 2 TW3 output transformers as chokes.

This snubber circuit is optional. It can help to reduce hum or hiss that some tubes may produce due to heater-to-cathode leakage.

Parts

One of the aims of this project is reproduceability. All parts are easy to find and inexpensive. Except for large-value capacitors, all parts are NOS soviet production, still available in Russia or Ukraine. Thus, the output transformers, TWZSH (aka TW-3SH), are available in Ukraine at electronic markets and so are 6f5p tubes. The power transformer is TAH6, rated at 56W, offering 2 6.3V, .9A windings and a 520V, 70mA, center-tapped coil. The rectifier is solid state - partly for simplicity, partly because TAH6 does not have enough filament power to drive a tube rectifier. Rectification is full-wave, performed by SF36 - super fast (50ns reverse impulse) low noise 3A, 400VRMS diodes. The PS filter is Pi-type, formed by a common 100uF electrolytic capacitor, and a 1.2k, 94uF filter for each channel. All resistors are carbon, 1/2 to 2 W. Signal capacitors are paper BMT. Paper & oil K40-y9 is another choice. Electrolytics are Samsung, rated at 400V. Rubicon capacitors is another and better choice.

Construction

The chassis is constructed from aluminum L-shaped extrusions. The front extrusion is the the “front panel”, hosting the double volume pot and two “mood” controls (driver bias pots). These pots also allow to adjust balance by +-3db. Behind that the input jacks are mounted, close to the tubes. Behind them two output transformers are located at 90 degrees to the power transformer. The speaker posts are placed immediately after the output transformers.

The rear extrusion was drilled to host power cord socket, power switch and fuse holder. The rectifier diodes are soldered directly onto the power transformer lugs.

The power supply is visibly detached from the signal circuit. The space between the power transformer and the output transformers can be occupied by paper and oil bypass, ultrapath or B+ filter caps - these are all possible upgrades.

Another way to use this space is to host a small filament transformer, providing current sufficient for a tube rectifier, which can be located in the space by the power transformer side.

Soldering is almost universally point-to-point and directly onto the tubes, with the exception of the bias resistors and bypass caps of the power stage which are mounted on bias cap switch. The channels are wired symmetrically. Each channel has its own ground bus, short in length, connected to the input socket and not connected to the aluminum chassis.

Most parts are soldered on the bus in close proximity (< 20mm) of these points thus forming “star grounding”. The power supply is also symmetrical (common rectifier, split filters) and has its own ground bus. The power supply bus is connected to the chassis and via separate wires - the grounds of each channel.

Two wooden blocks are attached to the side L extrusions. Those shown on the picture are prototypes. The final version will sport more exotic wood, with access to the binding posts cut through the wood. Also, missing on the pics are the top cosmetic covers The output transformers are supposed to be covered with metal boxes, and so is the power transformer. the remaining top surface can be covered with small anodized aluminum panels, ceramic, copper, nickel alloy, etc.

Measurements

We loaded the amp with 8 ohm, 2W resistors and obtained distortion data and frequency sweeps. The results are below. All data were obtained with 0db level corresponding to the output signal amplitude at which the amplifier starts clipping, which is the amplitude that gives raise to lots of high order harmonics. Unless specified, the output level is -3 db under the level of clipping.
First, the spectrum data for various settings of the bias, load and bypass controls. Some combinations offer interesting possibilities for tone experiments.

These patters are mirrored by the other channel as well.

The frequency response plot into 8 ohm resistive load, with the bypass cap, at full power is 0..-3dB in 30Hz-20kHz region; with bypass cap disconnected: 0..-2dB in 30Hz-20kHz region.

More detailed data is in [5]

Variants

This design was repeated, with some variations, by me and by friends. Here are a few picks.

San Francisco prototype

Viktor Dodatko, Dnepropetrovsk:

Orest Baidan, Mountain View:

http://www.geocities.com/dmitrynizh

I will instead concentrate on the more basic functionality of output transformers. Power transformers are pretty basic and the things that are necessary (read “basic knowledge.” There is more but not necessary to fully know to make a good power supply. This is covered in the power supplies section) to understand about them is turns ratio and power handling (for example, the voltages of the secondaries versus the current).

However, in audio circuits, a good understanding of the two types of output transformers is necessary to make a good choice for the best sound and efficiency.

WHAT IS A TRANSFORMER?

As the word suggests, a transformer transforms one thing in to another. In this case we are transforming one level of voltage and current to a different voltage and current. With that we are also transforming one impedance to another. Before I go on it is necessary to remember that transformers work with AC only. Nothing will happen if a constant DC is used. The AC induces a varying magnetic field into a coil of wire that also reinduces a current into the wire. This is known as self inductance. However, the coil can also induce a current into another coil.

A transformer is made using many windings of copper wire that is coated with acrylic to allow for insulation while allowing the wire to be as close as possible to itself in the winding. This only makes one coil of the transformer. One needs a second coil of wire with properties (number of windings) that make it convert the first coil’s characteristics. In other words, if the first coil has 5000 windings, and the second coil has 1000 windings, then it is said that the transformer has a turns ratio of 5 to 1 (5:1). That means that if I put an AC voltage of 100 volts on the first coil, also known as the primary, then the induced voltage on the second coil, also known as the secondary, when placed next to the primary will have a voltage induced to it that is 1/5th the size, or 20 volts AC. This is known as loose coupling.

The thing that allows transformers to conduct from one coil ot the other is the sharing of flux. This is known as mutual inductance. The air is the conductor of magnetic flux in this case. One can only get a certain amount of power this way. Air has a permeability (allows magnetic flux to conduct) of 1. Iron, on the other hand, has a permeability of about 1000 or more (steel goes up to about 5000). If I were to place a piece of iron in the center of the coil, then the lines of flux will be concentrated (because more magnetism can conduct easier through it, like the path of least resistance) and more power could be transferred to the secondary. What happens here is that more flux lines from both coils share the same path, or both coils have common lines of flux. This is what coupling means. If I make that iron core common to both, then more power can be transferred still.

Old style radios (in the 10’s and 20’s) used two separate coils connected by a “U” piece of iron as the common core. But there is an even more efficient configuration that is now used that also makes the transformer more compact. It is the practice of winding the secondary coil directly on top of the primary (although there are still some transformers that have two separate coils that use
a common core). This allows the lines of flux to cut through the secondary directly and more instantly. This is called unity coupling.

So now we know about turns ratio and its direct connection to voltage transfer. But in the voltage transfer there is an interesting thing that occurs. Current also changes. If I put that 100 volts into the transformer of 5:1 ratio the voltage drops to 20. But if that 100 volts came in as 1 amp, the secondary can provide up to 5 amps! This is the conservation of power rule that transformers must follow. Of course, this is not exact, since there are losses that occur in transformers, as will be discussed later. This applies to all transformers and there are of course a few more factors that come into play.

For example, a transformer rated at 12.5 volts output at 100 milliamps only means that the wire is thinner and there are many more windings. BUT the turns ratio is still maintained. So the primary of 125 volts only drains 10 milliamps, because the turns ratio is 10:1, so the input current is 1/10th of 100 milliamps.

IMPEDANCE IN A TRANSFORMER

There is a reflected impedance back to the primary that occurs from the secondary and the source, in this case speaker and the tube. That is why many say that there is a better damping from a triode than a pentode, because the transformer reflects the triode’s plate resistance A.K.A. its impedance. So we do not need to worry too much about that. However, the reflected impedance from a speaker as a load does vary nonetheless because speakers are not constant impedance loads and can vary from less than one ohm to more than 40 ohms. Ideally though the primary impedance is constant and is proportional to the turns ratio by the formula:

Zs/Zp = (Vs/Vp)^2

This states that the ratio of output impedance to input impedance is the same as the ratio of output voltage to input voltage squared. So, if the impedance ratio changes, the voltage ratio changes by a small amount.

For example, let’s use the common impedances of 5K for the primary and 8 ohms for the secondary of a single ended amplifier. Lets use 320 volts for the power supply. Now assuming typical values for the output tube, lets assume that 20 volts will be dropped across the tube/cathode resistor during normal full output swing. So we have 300 volts peak to peak, which translate to 150 volts peak. Multiplying 150 by .707 we get the RMS voltage at the primary. This comes out to 106.5 volts RMS. With this info we can substitute the variables and derive the secondary voltage with impedances as they are. (We will assume that the plate resistance is high enough not to affect the primary impedance too much so we use the 5K as is. Real world values will vary due to plate resistance and load). So we now plug in the known values to the formula and do some algebra to arrive at the secondary voltage (sqrt means square root):

8/5000 = (Vs/106.5)²

sqrt(8/5000)=Vs/106.5

[sqrt(8/5000)] X 106.5 = Vs

0.04 X 106.5 = Vs

4.26 = Vs

Now that we have the secondary voltage out we can check this by plugging all the now known values to the
original formula:

8/5000 = (4.26/106.5)²

0.0016 =  (0.04)²

0.0016 = 0.0016

From this we can see that if a four ohm speaker were put into the eight ohm output the primary impedance will change by the proportions involved. Lets see how that works, assuming that the voltage ratio’s don’t change, since the voltage ratio is directly proportional to turns ratio.

So we have the known values 4 ohms, 106.5 volts and 4.26 volts (this gets hairy!):

4/Zp = (4.26/106.5)²

sqrt(4)/ sqrt(Zp) = (4.26/106.5)

sqrt(4) = (4.26/106.5) (sqrt(Zp))

sqrt(4)/(4.26/106.5) = sqrt(Zp)

2/0.04 = sqrt(Zp)

50 = sqrt (Zp)

2500 = Zp

So, interestingly it seems that the proportion one would expect occurs when the impedance of the secondary changes. A direct proportion occurs. Going from eight ohms to four ohms (halving) actually lowers the primary impedance by the same factor, namely in half. This keeps the power transfer the same, because halving the primary impedance causes the effective current to double. But reflected impedance could still go as low as 625 ohms if the speaker impedance goes down to 1 ohm for a given frequency.

This is pretty much true for solid state also. Since the voltage levels are constant, the current is what changes. Good old Ohms law!

PARALLELING TUBES

So now, what if the opposite were true? What if I double the output tubes? Some have argued that doubling the output tubes have no effect on output power. Let’s take a look at this mathematically.

Let’s assume a plate resistance of 22500 (for a 6l6GC. My current pet output tube) in parallel we use the ole parallel formula. Actually it’s easy because it is merely half for two tubes. So we get about 11250. In parallel with the primary, though we get a different result. This is the formula for paralleling two unlike
values:

R1XR2/R1+r2

11250X5000/11250+5000

5625000/16250

3461

The Primary impedance now becomes 3461 ohms. So the reflected output impedance is about 5.2 ohms. Now, judging from this, what is the power? The voltage still remains at 106.5, so using a little ohms law we get a current of 106.5/3461, or 30.7 milliamps. Not a lot, is it. 106.5 times 30.7 milliamps is 3.2 watts. Not much. Let’s
see what it was originally.

With the above assumptions, namely the 6L6 (22500 ohm plate resistance) and a 5000 ohm primary we use the parallel formula and get a primary impedance of 4090 ohms! Pass 106.5 across this and we get 26 milliamps! So we have a power output of 2.7 watts!
Not a doubling of power, but an increase of half a watt, but an increase none the less.

However, it is noteworthy to add that I am calculating these levels based on an unbypassed cathode resistor in a cathode biased amp. Bypassing the cathode resistor decreases the plate
resistance, hence impedance.

So this puts to rest the argument that double the tubes gives double the power. However, in order to take advantage of the doubling of tubes, one needs to half the input or primary impedance of the transformer. Since power through a transformer must be identical on both sides (barring intrinsic losses), this is possible without increasing voltage. So assuming that both tubes combined allows 30.7 milliamps through at 106.5 volts each, the output power will be 6.4 watts. But this means that the primary impedance must be 2500 ohms. The turns ratio will be different, allowing for the current ratio. Current ratio is inversely proportional to turns ratio. In other words:

Np / Ns = Vp / Vs = Is / Ip

Where Np and Ns are the turns in the primary and the secondary respectively, Vp and Vs are voltages and Ip and Is are the currents. If we really want to go crazy, we could calculate this all the way (impedances, voltages, etc.), but I think you can use this as a practice example for yourself.

TRANSFORMER LOSSES

As I said before I will leave core and magnetism theory to the textbooks. But I will mention that the core materials, winding practice, and wire quality all come into play where sound quality is concerned. That is why there is (relatively) so much of a variety in manufacturers and within manufacturers as to the transformers cost and type.

One of the losses that occurs in a transformer is known as eddy currents. What happens is that the current that flows through the coils induces magnetic currents within the core material. This in turn induces a small current back into the coil. It is very similar to self induction. This however causes two of the problems. One is to lower the current transferred to the secondary, so this is a current loss, and it generates heat within the transformer. This is heat loss. This is one of the reasons why the transfer efficiency is 80-90%.

Another loss is stray capacitance. As with any conductor, if it runs close to another conductor in parallel, they will have a capacitance between the two. A transformers windings are many parallel wires. Then there is capacitance from primary to secondary, and from both to the core, which in most cases connected to the chassis, or ground. This limits high frequency response.

There is also a phenomenon called hysteresis. This is a delay that is caused by the time it takes for the core to magnetize and release its magnetism. Different formulations of iron or steel have different degrees of hysteresis. Unfortunately the formulations that have less hysteresis also do not perform as well. So a balance must be maintained. Hysteresis causes phase distortion in the low frequencies. This is not the same as the hysteresis used in some circuits to speed up their switching time. That is a form of positive feedback.

SINGLE ENDED AND PUSH-PULL TRANSFORMERS

The single ended transformer is an interesting beast. Talk about compromises! There needs to be a balance of power and frequency response here. Make a single ended transformer produce a full range, and one needs to sacrifice some power. Make an SE transformer for power and frequency response suffers. There have been some very good posts on the news group rec.audio.tubes about the technical aspects of this relationship.

The frequency response of an SE (low frequency) depends on the primary winding inductance. In order to get a large inductance one needs to add windings. This can lead to saturation easier, though. It also leads to a higher impedance and lower current. Lower power. To lower the ill effects, less windings are needed, but this reduces low frequency output. What a dilemma. So we need to focus on a happy balance.

In the SE design, one needs to take into consideration the saturation point of the transformer. If one biases the output tube past the midpoint, then they run the risk of saturating the core, causing distortion, similar to clipping. The reason for the precaution it because there is always a single direction DC current making the transformer into a single magnet. I am using the word single because in a push pull transformer there are two flows of current, but more on that later. So in essence we have a fluctuation of magnetism from midpoint to full to none, which corresponds to the midpoint flow of current, to full current to none. The same is the case with voltage. The saturation problem is minimized by several techniques. There is winding, core material, and a small gap. This gap is put between the I bar and the E portion of the transformer. (Fig. 1)

Then there is the reduction or elimination of the sweet sounding even order harmonic distortion. I have yet to find out why this occurs, except to surmise that the even order components are in phase, while the odd order components are out of phase. So the even order components cancel out in push pull while the odd order reinforce each other. I think in the strictest sense that the real effect has to do with the speed that the components reach saturation and cut-off as opposed to pure Fourier analysis, but that is discussed on the distortion page.

Here is a couple of posts by Mike LaFevre of Acrosound about some particulars of transformers. I think it is pretty interesting and in depth.

CONCLUSION

So, to SE or not to SE, that is the question. I personally prefer push pull, but I have made a SE amp and find it to have some desirable sonic qualities as well as surprising power.

There are quite a few manufacturers of quality power and output transformers available. Here are a few manufacturers and vendor of them. Note: they are in no particular order:

Manufacturers:

One Electron
Lundahl
Magnequest
Hammond

Vendors:

Antique Electronic Supply
Triode Electronics Corporation
Acrosound
Communication Jute

Designing a SE output transformer is the imposition of the transformer’s gods’ fury…. look at the formula’s for AC flux density vis-a-vis DC flux density. What we find is that if we want to keep the AC flux density low then (all other things equal) we would want a large number of primary turns. But, conversely, if we want the DC flux to be low (all other things being equal) we would want to decrease the number of primary turns.

In part of the design process for SE… you must calculate what your AC flux Consumption will be (at full power and lowest frequency of interest) and then subtract this amount from the maximum flux capacity for the specific core material that you are using. The amount left over tells you what the maximum DC flux could be…. if you wish to stay below the knee. And this, provided only as a one approach to design, assumes that your willing (or feel it smart) to use all of the flux available. And there are good arguments against this.

Another approach as explained in detail in Rueben Lee’s book on magnetics is that to maintain a high degree of inductive consistency… then your DC flux should be(I think it was… short of looking up the reference) 3 to 4 times the magnitude of your AC flux. And remember in a magnetic component with both an AC and a DC flux component the effective perm is a function of the absolute values of each as well as the relative (to each other)ratio of fluxes. As just one example… on one of our part’s (NLA in the DIY community btw) the DC flux was approximately 9850 gauss while the AC flux was approximately 3000 gauss. Rueben Lee demonstrates that if these conditions hold then the deviance in primary induction will be approximately 20% or so.

Push-pull coils can (and perhaps should be) mirror-imaged around a centerline (both geometric and electrical)… so that each half has the same leakage inductance and each half has the same capacity. And given the “equal but opposite” (at least in theory)nature of the two halves of a PP transformer the coil design and it’s geometry will be (or do I mean should be?) different than a SE output transformer design would be.

The SE output has one end of it’s winding at AC ground potential and the other end at “AC HIGH”…. for push-pull we have two ends at “AC HIGH” and (let us say informally) the center (the B+) is at AC ground. This is a fundamental difference…. if I am making this clear enough in my writing….. so if we follow the AC voltage gradients in a PP transformer we will find that they are “equal but opposite” in a mirror imaged way between the two halves of the coil.

SE outputs don’t offer us this “degree of symmetry” (or at least not on first blush)…. here one end of the winding is at AC ground and the other end at AC HIGH. So that as we traverse the coil, through each primary winding turn, the AC voltage potential is changing… at our interstices (junctions between primary and secondary windings) we then have certain voltage gradients… the gradients will be different across each interstice… and the capacities different. So if you have a PP output that was optimized for a different voltage gradient and a different voltage potential at the interstices… then using it as a SE device the unit will not achieve the intended optimization that the transformer designer had wished to achieve.

There is a way to do a SE output transformer as a quasi-symmetrical design…. wherein you design it as a PP coil geometry but then reverse wind one half of the primary and put it in parallel with the other half. Using this technique a design can achieve the same level of “symmetry” in terms of the capacities being equal about a coil geometric centerline….

POST 2

>Mike, please excuse my technical misunderstandings in advance. In your
>post, considering symmetric winding geometries for SE transformers, you
>described a technique of taking two primary windings, as used in a PP,
>reversing one winding, and connecting them in parallel. Question: are
>the two windings not bucking each other, if paralleled out of phase?
>If an ordinary PP transformer was reconfigured this way, would DC
>currents in the paralleled windings be nulled out?

Several of the responses to my post thought that the example used a bilfilar wound primary winding. Although perhaps it could, the illustration does not rest on it being bifilar wound. And the language I used was imprecise enough to lead to this confusion.

In the example…. what is happening is that you wind the whole primary twice… using three guages smaller than normal. for instance suppose the primary needed 2,000 turns of #30. then what you would do it wind 2.000 turns of #33 in P1 and P3 and another 2000 turns in P2 and P4. The example I had in mind derives from the Acrosound patent… and is a modification
of that patent in that P2 and P4 are reverse wound and then placed in parallel with P1 and P3.

Barry concerns about phasing… remember, and I always have to think this through and on some coils with tons more interleaving it becomes a real puzzle to keep the reverses straight…. when you reverse a winding the physical start becomes the electrical finish. And the reverse physical finish becomes an electrical start. so if you go through these reverses…. and their placement in parallel with the non reversed windings (say P1 an P3) then the electrical
polarities of the reversed windings… you’ll see (I hope) that they are not bucking….

>Question: Can an SE
>winding symmetrically configured give true symmetrical cancellation of
>stray reactances, as reflected to the secondary? Its not clear to me how
>symmetrical, out of phase components could be set up in an SE winding,
>in order to effect a true cancellation. I hope there’s a cogent question
>in here.

I used the term quasi-symmetrical moreso as an analogy to describe the physical configuration of the windings and their electrical polarities in response to a question about the capacitances within a coil and etc. In a single ended output…your right…. no differential phases (the push and the pull) are present…

So the strategy in illustrating what I called a quasi-symmetrical coil geometry in a SE output is to achieve a minimum number of different voltage gradients across the interstices of primary to secondary and to achieve a “sameness” of voltage potentials across these interstices. Wish you (and everyone who is interested in this topic) had a copy of the Acrosound patent and perhaps a diagram showing the winding sequences and etc. Then it would be easier to explain…

Just another note…. the illustrations of coil geometries should not be construed (on the basis of my choosing it as an illustration) as the method that I use on our own products. I actually use several different strategies for interleaving in our own products… but to fend off any misconceptions or speculations… I just wanted to put this disclaimer in here….

POST 3

I have been threatening Joe Roberts to do some more “Core Issues” in Sound Practices for quite some time and this is one of the articles that I would like to do.

Anyone have access to the old SP where I did an article on “how to pick a power transformer”? In it there was a discussion of current capability and how to evaluate or, minimally, establish a context for assessing qualitatively the numbers (the current ratings) which mfgr’s spec out.

Part of the problem in power trans is that the “current” number must be provided in some sort of context if it is to have meaning as a useful basis of discriminating btwn products. My article tried to show that to compare “raw numbers” from one company against “raw numbers” from another was often like comparing “apples to oranges” as opposed to “apples vs. apples”. My sense is that the same difficulty applies to especially the comparison of raw DC current numbers which manufacturers use for SE output transformers. But…. least I get a head of myself…

Steve, I don’t mind putting up some of my ideas and research on
“AC versus DC characteristics in SE and PP” as you stated above…. but I want to figure out how to structure the response so that it goes in some logical order. And I would be grateful for constructive comments and criticisms since I want to turn some of this stuff (that I have done over the past week or so) into an article for Sound Practices or Positive Feedback.

so give me a day or two or three or four or….. yep…. probably never get it done… so let me throw out a teaser and see if any of you RAT’s can “tell” me about this formula and what it means and what the variables are and what is unique about some of the variables… it is one of the building blocks of designing a SE output trans…. a focus and understanding of it…. leads, IMO, to clearer sight on some of the other issues that come after it.

energy factor equals L times (Isubdc) squared divided by core volume

core volume equals L times (Isubdc) squared divided by energy factor

L is primary inductance — unit measure is henries

Isub dc is the magnitude of dc current (and it is squared) — unit of measure is ADC

core volume is the volume of the magnetic core being used — unit of measure is cubic centimeters

Excerpt from Rec.Audio.Tubes newsgroup
By Mike LaFevre

Fig. 1

Fig. 2

In the signal domain a triodes cascade stage is reproduced in Fig.2.This schematic can represent most of the circuits that you can meet really, Figg. 3-6.

Fig. 3

Fig. 4

Clearly, other variants exist with pentodes and/or buffers that can be represented with the scheme of Fig. 2. These circuits in their disarming semplicity have a good fitting with the HCT, further they result easy to build and above all are good sounding. The variants of this schemes in wich the coupling between the stages happens with an iron (inductor or transformer), Fig. 5-6, are more immediate and easy to manipulate since it’s often sufficient to alter the resistor in the grid’s output tube to put the HCT in the best condition of operation without altering dangerously the bias of the drive stage. On the power amplification side, the only constraint is, in my opinion, the accurate choice of the tube for the drive stage. The main characteristics

Fig. 5

Fig. 6

are a “good” m and a low rp (that is an high gm) as well as a meaningful power dissipation. Canonical receiving tubes used in traditional amplifier design, like ECC83/12AX7, 6SN7 and so on, can result unsatisfactory therefore I prefer vacuum tubes with best electrical performances like WE437A, 3A167M, E55L, E810F and others. Unfortunately the latters are rarer and expensive. Of course, you can resort to composite configurations like SRPP, MU-Follower’s, and Cascode+Cathode_Follower in order to exceed the inherent limitations of standard tubes. I stress the importance of a good driver because they result mandatory if you decide to design a class A2 amplifier or you take aim at exploring the enormous potentialities of direct heating big transmitting tubes like 211 and 845 that usually can present grid current already for weakly negative grid voltages. Further in this tubes the input capacities are not negligible and a wide driving signal is requested.

From Fig.2, you have:

Ip1=g11*Vin1 + g12*Vin1^2 (1); Ip2=g21*Vin2 + g22*Vin2^2 (2);

where gij coefficients are costants inherent to the mutal dynamic charactersitcs of the two gain stages. You can extracts these coefficients with a mathematical or experimental method. Within the mathematical approach you can follow an analytical or numerical way. I prefer the latter because it connects speed and good precision without the idiosyncrasies of the simbolic calculus. For triodes, assuming that the effects superior to the second order are negligible, and observing that:

Vin2=-Z1*Ip1 (3); Vout=-Z2*Ip2 (4);

by replacing the (1) and (2) in (3) and (4) rispectively it’s easy to reach at the following expression for the output voltage:

Vout= g11*g21*Z1*Z2*Vin1+(g12*g21*Z1*Z2g11^2*g21*Z1^2)*Vin1^2+2*g11*g12*g22*Z1^2*Z2*Vin^3 (5)

where:

Vin1=Vp*sinwt (6)

and therefore:

Vin1^2= 1/2*Vp^2*(1-cos2wt) (7) Vin1^3= 1/4*Vp^3*(3*sinwt-sin3wt) (8)

(6) is the mathematical expression of the input signal while (7) and (8) are derived with a bit of trigonometry.

By observing (5) two main considerations emerge:

· Also in the hypothesis of a 2nd order mutual dynamic characteristic for the common cathode stages the output signal present a 3rd order component;

· The algebraic expression between round brackets underlines the theoretical possibility for an annulment of the 2nd order component and thanks to (7) also for the rectified component; Unfortunately the exacts values for Z1 and Z2 that permits a reduction or nulling for the 2nd order effect it’s hard to find since:

(g11,g12)=f(Z1) (9) (g21,g22)=f(Z2) (10)

(where d1, the “error” produced by the first triode, includes distortions, noise, hum, drift and so on); next the criterion network b (that could also contain reactive element for frequency compensation) attenuates the amplified signal by the value

b = 1/G1 (11)

and finally:

Vin1= -G1*Vin + d1 (12) Vout= -G2*{[(-G1Vin + d 1)*b] + d2} (13)

where likewise the first stage G2 and d2 are the gain and the error for the second one rispectively. By assuming:

G1= G2= G= 1/b; (14)

eq. (13) is reduced to the following:

Vout= G * Vin - d1 + d2 (15)

and d1 and d2 are similar if:

a) both triodes are similar in their dynamic behaviour;

b) the input signal swing it’s small;

Fig. 7

therefore a lower value in the output signal THD can result. The reduction mechanism of even harmonics don’t have clearly the same effectiveness that you can find in ideal push-pull structures, where the input signals are really equal altough out-of-phase. In cascade stages as in Fig. 7 the 2nd stage receive and out-phase signal with the “harmonic imprint” caused from the 1st one. I have defined an Improvement Index (II):

(16)

and its graphic shows good performances of the cascade for small input signals, where the harmonic content in the 1st stage output is not elevated, Fig. 8. For both tubes and bias points please refer to [2].

Fig. 8

The mechanism that characterizes the behaviour of the circuit in Fig.7 could be used, duly adapted, for the realization of an electronic phase inverter for push-pull amplifiers and for a line stage with low THD, Fig. 9. A similar preamplifier presents a cumulative gain as the first stage, a low THD and, thanks to cathode-follower, a low output impedance.

Fig. 9

Clearly, other circumstances could require different bias points and/or vacuum tubes for the basic structure in Figs. 7,9. For this situations an “ad hoc” examen permits the best application of the HCT although the procedural mechanism is the same.

References:

1 Radiotron Designer’s Handbook RCA, pp. 509, 580-81, Chap XIV, 1953
2 RCA Receiving Tube Manual Technical Series RC-19, p. 344 1958